Ultrasonic fault detection system using a high dynamic range analog to digital conversion system

ABSTRACT

A method and apparatus for effecting ultrasonic flaw detection of an object processes an echo signal received from the object being tested in at least three signal channels, wherein the echo signal is scaled to different degrees along each channel to increase and extend the dynamic range of an associated A/D converter system, in a manner which dispenses with the need for using numerous analog high pass and low pass filters and a variable gain amplifier. This reduces complexity and avoids performance limitations. The digital to analog converters sample the differently scaled input signal and a selection circuit selects the output of the digital output obtained from that analog to digital converter which has the highest gain, but which has not overflowed. The digital outputs are seamlessly merged to produce an output that can be displayed as a scan display which shows the location of faults.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application claims the benefit and priority of U.S. patentapplication Ser. No. 11/489,889 filed Jul. 20, 2006 entitled ULTRASONICFAULT DETECTION SYSTEM USING A HIGH DYNAMIC RANGE ANALOG TO DIGITALCONVERSION SYSTEM and U.S. Provisional patent application Ser. No.60/726,798 filed Oct. 14, 2005 entitled ULTRASONIC FAULT DETECTIONSYSTEM USING A HIGH DYNAMIC RANGE ANALOG TO DIGITAL CONVERSION SYSTEMand U.S. Provisional patent application Ser. No. 60/726,776, filed Oct.14, 2005 entitled ULTRASONIC DETECTION MEASUREMENT SYSTEM USING ATUNABLE DIGITAL FILTER WITH 4× INTERPOLATOR, and U.S. Provisional patentapplication Ser. No. 60/726,575, filed Oct. 14, 2005 entitled DIGITALTIME VARIABLE AMPLIFIER FOR NON-DESTRUCTIVE TEST INSTRUMENT, the entiredisclosures of which are incorporated herein by reference.

BACKGROUND OF THE INVENTION

The present invention generally relates to ultrasonic inspection systemsutilized to detect internal structural faults, e.g. cracks,discontinuities, corrosion or thickness variations within an object or amaterial, for example, in such crucial structures as airline wings. Thisis done by transmitting ultrasonic pulses to a target object andanalyzing echo signals detected from the target object. Moreparticularly, the present invention relates to a high dynamic rangeanalog to digital conversion system and method which can be used in suchultrasonic inspection systems, particularly whereby the object isscanned with an ultrasonic probe or transducer. The present inventionalso relates to eddy current inspection systems utilized to detectinternal structural faults.

The prior art of ultrasonic flaw detectors is exemplified by suchproducts as the instant assignee's Epoch 4 Plus product. Competitiveproducts available from General Electric are known as the USM 35×, USN58L and USN 60 fault detection systems. In general, prior art ultrasonicflaw detectors utilize highly complex analog front ends that containmany parts which pose especially difficult problems in terms ofcalibration, reliability, set up time, consistency of results andoptimization for specific usages and settings.

Typical prior art ultrasonic flaw detectors include a transducer whichis placed against the object to be tested and which works in conjunctionwith numerous analog circuits such as gain calibrators, preamplifiersand attenuators, variable gain amplifiers, and high pass and low passanalog filters that operate over many different frequency bands andwhich need to be carefully calibrated and maintained.

As a result, present flaw detectors present a host of problems todesigners and users of such equipment, which impact theirtroubleshooting and repair owing to their complexity. These problemsinclude such issues as matching input impedances seen by the transducerwhich changes with different gain amplifiers that are switched in andout of the signal path. This adversely impacts the frequency responseand introduces various gain nonlinearities. It poses issues ofcalibration, as analog circuits are switched in and out of the signalpath.

Another problem with existing flaw detectors is attributable to theirback wall attenuation performance which impacts the ability to detectflaws that are located very near the back wall of the object beingtested. This problem poses particular problems with the time varied gainfunction which has a limited gain range and gain rate of change in priorart devices.

Another prior art drawback ensues from the manner in which analogcircuits are coupled, which results in each operational amplifier in thesignal path having different DC offset errors that must be nulled inorder to keep the input signal at the mid-point of the analog to digitalconverter being utilized, in order to allow the maximum full amplitudescale of such converter to be utilized. Furthermore, the DC offseterrors can cause the waveform presented on the display to not becentered vertically on the waveform portion of the screen, therebycausing an undesirable anomaly in the waveform that the operatoranalyzes to determine the results of their inspection. The error nullingprocesses in the prior art are therefore unreliable, particularly athigh gain, due to DC baseline measurement inaccuracies caused by noise.

The intensely analog implementation of the front ends of existing flawdetectors poses further issues owing to the need to utilize the entiredynamic range of the instrument to be utilized which creates variousgain linearity calibration issues.

An ultrasonic inspection apparatus of the prior art is described in U.S.Pat. No. 5,671,154, which provides background information for theapparatus and method of the present invention.

SUMMARY OF THE INVENTION

Generally, it is an object of the present invention to provide anapparatus and method for ultrasonic inspection of objects which avoid orameliorate the aforementioned drawbacks of the prior art.

It is a further object of the invention to provide an ultrasonicinspection apparatus and method that is implemented in simplercircuitry.

It is a further object of the present invention to provide an ultrasonicinspection apparatus and method that requires a shorter and simplerprocess of calibration and adjustment prior to use.

Yet another object of the invention is to provide an ultrasonicinspection apparatus and method that provides an electronic inspectionapparatus and method that delivers more accurate and more easilyreadable and consistent inspection results.

The foregoing and other objects of the invention are realized by amethod and apparatus that extend the dynamic range of the A/D convertercircuit and eliminate the need for a Variable Gain Amplifier (VGA)circuit and its associated complexity and performance limitations.

According to one aspect of the invention, the apparatus and method ofthe invention are embodied as a multiple A/D circuit that includesmultiple channels coupled to receive a single analog input signal, eachof the channels having the means for converting the analog input signalto a digital signal.

Another aspect of the invention includes: a means to adjust therespective sample times to compensate for all sources of timing skew,including the propagation delays of each preamplifier and any othersource of skew revealed by examination of the A/D converter output data;a means for preventing saturation of the input stage of each channel'spreamplifier to prevent signal distortion from affecting the inputs tothe other channels; a means for adjusting the frequency response of eachchannel to substantially match, as well as adjusting the overallfrequency response of the apparatus; a means for detecting a channeloverflow condition in one or more of the channels having higher gain;and a means for merging the multiple channels into a continuous outputstream.

According to another aspect of the invention, the multi-channelconverter circuit of the invention includes a means for eliminatingsignal offset errors in each channel by injecting DC signals from a D/Aconverter at various points of the analog signal path to null out theoffset errors.

According to another aspect of the invention, the means for merging themultiple channels is operable as a function of a result generated by thechannel overflow condition detection means. Furthermore, the means formerging the multiple channels is operable to output a result of thechannel having lower gain when a channel overflow condition is detectedon any of the channels having higher gain.

According to another aspect of the invention, the means forsubstantially matching the frequency response of each analog channel isprovided to minimize amplitude matching errors between channels,particularly at high frequencies.

According to another aspect of the invention, each A/D converter circuitincludes a means for changing the reference voltage to adjust the fullscale range by using a D/A converter. This is used to optimize signalamplitude matching.

According to another aspect of the invention, the multiple A/D convertercircuit of the invention includes a means for matching the result ofeach channel with different gains.

According to another aspect of the invention, the multiple A/D circuitof the invention also includes a means for adjusting placement of therising edge of the sample clock of one channel in time with respect to aclock circuit portion of another channel so that the sample times ofeach channel are adjusted to compensate for the propagation delays ofeach preamplifier channel and any other source of skew revealed byexamination of the A/D converter output data.

According to another aspect of the invention, the channel overflowcondition detecting means further comprises a means to extend the timeduration of the overflow signal that comes from the A/D converters inorder to ensure that all amplifiers in the signal path from the firstamplifier to the amplifiers internal to the A/D converters have adequatetime to return to their linear region of operation.

According to still another aspect of the invention, the means formerging multiple channels further comprises a means for adjusting, e.g.scaling, a data bit position of a result of a channel, or channels,having lower gain to match a result of the channel, or channels, havinghigher gain. This can be accomplished, for example, by bit shiftingusing a shift register, a multiplexer and the like, or by any means.

According to yet other aspect of the invention, methods for convertingan analog signal to a digital signal are provided which include, forexample, splitting an input analog signal into larger and smaller signalchannels; scaling the input signal on the larger and smaller signalchannels such that the smaller signal channels have higher resolutionthan the larger signal channels; sampling the larger and smaller signalchannels using separate A/D converters; and outputting the result of oneof the larger and smaller signal channels as a function of determiningwhether the larger signal channel is valid.

The methods of the invention also include merging the results of thelarger signal channels with the results of the smaller signal channelsinto a merged result; and outputting the merged results.

Other features and advantages of the present invention will becomeapparent from the following description of the invention which refers tothe accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is block diagram of a basic arrangement of an ultrasonicinspection apparatus.

FIG. 2 is a basic waveform diagram for the device of FIG. 1.

FIG. 3 is a waveform diagram illustrating the trailing edgecharacteristic of an ultrasonic pulse.

FIG. 4 is a block diagram that provides a side-by-side comparison of awaveform display with fault locations in a target object.

FIG. 5 is a continuation of FIG. 4.

FIG. 6 illustrates a circuit block diagram of a prior art implementationof an ultrasonic inspection apparatus.

FIG. 7 is a circuit diagram of a digitally intensive implementation ofan ultrasonic inspection apparatus in accordance with the presentinvention.

FIGS. 8 a and 8 b are further block diagrams of a further implementationof the present invention.

FIG. 8 c corresponds to FIG. 8 b, but includes purely digital DC offsetcompensation.

FIGS. 8 d and 8 e correspond to FIG. 8 b, but utilize magnitudecomparators instead of overflow indicators, with FIG. 8 e adding digitalbase line correction.

FIGS. 8 f and 8 g correspond to FIG. 8 b, but add base line correctionin each channel.

FIG. 8 h corresponds to FIG. 8 b, but includes delay circuits fordealing with fast slewing input signals.

FIG. 9 illustrates a circuit block diagram of an alternate embodimentfor the Front End section delineated in FIG. 7.

FIG. 10 is a signal diagram utilized to explain certain conceptsapplicable to the operation of the circuit in FIGS. 8 d, 8 e and 8 h.

FIG. 11 is a block diagram of a blending circuit associated with FIG. 8d.

DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS OF THE INVENTION

Reference is initially made to FIGS. 1 and 2, to provide backgroundinformation on the general environment of and various problems solved bythe present invention.

In FIG. 1, an ultrasonic transmit-receive unit 10 transmits anelectrical pulse signal 10 a at a predetermined period to a probe ortransducer 12 which is coupled to a target object 14, such as to steelmaterial, directly or through a delay material such as water or quartz.As shown in FIG. 2, the probe 12 converts the trigger pulse signal 12 ainto an ultrasonic pulse 10 a which it transmits through the targetobject 14. The ultrasonic pulse 10 a applied into the target object 14is subsequently reflected by a bottom surface 14 a of the target object14 and received by the probe 12. The probe 12 converts the reflectedwave into an electrical signal which is supplied as an electrical echosignal 10 b to the ultrasonic transmit-receive unit 10. The ultrasonictransmit-receive unit 10 amplifies the electrical signal 10 b andtransmits the amplified signal 11 to a signal processing device 16 as anecho signal 11. As used herein, the term probe or transducer includesembodiments which implement the transducer using distinct transmitter(s)and receiver(s).

The echo signal 11 includes a bottom surface echo 11 a corresponding tothe wave reflected by the bottom surface 14 a and a flaw echo 11 bcaused by a flaw 14 b in the object 14. In addition, the frequency ofthe ultrasonic echo pulse 11 is determined primarily by the thickness orother property of the ultrasonic vibrator incorporated in the probe 12.The frequency of the ultrasonic pulse 10 a used for inspection is set totens of kHz to tens of MHz. Therefore, the frequency range of the signalwaveforms of the bottom surface echo 11 a, and the flaw echo 11 bincluded in the echo signal 11 cover a wide range of from about 50 KHzto tens of MHz.

The signal processing device 16 performs various signal processing ofthe echo signal 11 received from the ultrasonic transmit-receive unit10, and the signal processing device 16 displays on a display unit 18,an output result that represents the presence/absence of a flaw or flawsand in some cases, the thickness of the target object 14. In order tosignal process the echo signal 11 and display the echo signal, a triggersignal S synchronized with the pulse signal 10 a is supplied from theultrasonic transmit-receive unit 10 to the signal processing device 16.

In the flaw inspection apparatus arranged as described above, the echosignal 11 output from the ultrasonic transmit-receive unit 10 includes,in addition to the bottom surface echo 11 a and flaw echo 11 b, acertain amount of noise. When the amount of noise included in theultrasonic pulse 11 is large, the reliability of an inspection result isconsiderably degraded. The noise is roughly classified into electricalnoise and material noise.

The electrical noise comprises external noise caused by mixing anelectromagnetic or electrostatic wave into the probe 12, the ultrasonictransmit-receive unit 10, connection cables, e.g., cables 13, or thelike, and internal noise generated by amplifier(s) and the likeincorporated in the ultrasonic transmit-receive unit 10.

Reduction of the noise included in the echo signal 11 is very importantto perform ultrasonic inspection at high accuracy. Conventionally, ananalog filter is used to reduce noise components included in the echosignal 11. For example, a BPF (Band pass Filter) is used to pass thefrequency component of the ultrasonic echo relative to the electricalnoise having a wide-frequency component. In addition, an LPF (Low-PassFilter) or a BPF is used for material noise, recognizing that thefrequency distribution of the flaw echo 11 b (FIG. 2) is lower than thatof the echo produced by signal scattering. In this manner, when ananalog filter is used, noise components included in the echo signal 11 bcan be reduced to a level equal to or lower than a predetermined level.

It is generally known that the frequency distribution of a flaw echosignal changes based on the ultrasonic attenuation characteristics ofthe target object 14. Therefore, when a BPF is to be used for materialnoise represented by a scattered echo or the like, a filter havingoptimal characteristics is desirably used in accordance with the targetobject 14. However, since the passing frequency characteristic of theanalog filter cannot be easily changed, a larger number of filters,having different passing frequency characteristics corresponding to thedifferent ultrasonic attenuation characteristics of the variousmaterials associated with target objects 14 must be prepared. In thismanner, when different filters are used in accordance with the materialcharacteristics of target object 14, practical difficulties occur inconsideration of operability or economic advantages versus the cost andcomplexity of the overall system.

In some cases, the flaw echo 11 b may be very close to the front surface14 c of target object 14 which will place it in close proximity to thetrailing edge of transmitted pulse 10 a. For this reason, it isdesirable for the end of the trailing edge (magnified as trailing edge10 at in FIG. 3) of the transmitted pulse 10 a to settle to the zerobase line 10 ab as quickly as possible in order not to interfere withthe returning flaw echo 11 b. The settling time 7 a to the zero baseline is a determining factor of a flaw detector's near surfaceresolution.

Considering that the gain of the ultrasonic transmit-receive unit 10 canbe adjusted up to 110 dB (as required by European standard EN 12668-1),a small amount of base line error prior to a gain amplification stage inthe ultrasonic transmit-receive unit 10 will cause a large error at theoutput of the gain amplification stage if the gain level is set toohigh.

The resulting base line error at the input to the signal processingdevice 16 will either:

(a) cause the dynamic range to be reduced because the maximum verticaldisplacement of the signal on the screen will be reduced by the amountof offset of the base line, which produces a reduction in theinstrument's sensitivity to detecting flaw echoes, or

(b) if sufficiently high in amplitude, cause a gain amplification stage,or gain amplification stages, to saturate, thereby preventing an echosignal from being detected at all.

Conventionally, the base line error problem described above is addressedin one of two ways. In accordance with a first approach, a HPF is usedin the signal path of the input of ultrasonic transmit-receive unit 10in order to filter out the low frequency content of the trailing edge 10at of the transmitted pulse 10 a. The trailing edge 10 at of thetransmitted pulse 10 a can be improved by the HPF as is indicated by theapproximated dotted line 7 c.

However, the effectiveness of the HPF solution is limited in severalmanners. First, the HPF cutoff frequency (f HPF −3 dB) must be as highas possible to minimize the low frequency content of the trailing edge10 at of the transmitted pulse 10 a. For example, if the excitationfrequency of probe 12 is 10 MHz and the f HPF −3 dB is 5 MHz, theundesirable effect on the receiver base line is greatly reduced.

Unfortunately, it is not uncommon to use an excitation frequency forprobe 12 as low as 500 kHz which would require the f HPF −3 dB to bebelow 500 kHz. The HPF solution loses much of its effectiveness in thisfrequency range because an undesirable amount of the low frequencycontent of the trailing edge 10 at of the transmitted pulse 10 a isallowed to pass through the HPF and contribute to base line error.

Secondly, the maximum amplitude of the transmitted pulse applied to afirst amplifier stage (not shown) of ultrasonic transmit-receive unit 10is limited (clamped) to a few volts in order to prevent damage to theamplifier circuit. It is common to operate the gain of the ultrasonictransmit-receive unit 10 at a level that will cause the amplifiers tosaturate every time the pulser is fired. If the filters are notcritically damped, the filter response after coming out of saturationwill cause the trailing edge of the transmitted pulse 10 a to be worsethan if no filtering was applied. It is possible for each manufacturedinstrument to have the numerous filters tuned to ensure criticaldamping; however, practical difficulties occur in consideration ofmanufacturability and temperature drift of the filter components.

It should also be noted that once an amplifier goes into saturation, ittakes a significant amount of time for the amplifier to return to thelinear region of operation. This causes the trailing edge of thetransmitted pulse 10 a to take more time to return to the zero base linethan would be the case if the amplifier input signal remained below thesaturation level (i.e. within the linear range of operation).

An alternate method used to address the base line error problem is todirectly couple the clamped transmitted pulse 10 a to the input ofultrasonic transmit-receive unit 10. This method avoids one of theproblems described above, because no HPF or BPF filters are used.

The effectiveness of the direct coupling solution is limited in twoways. First, it does nothing to reduce the low frequency content of thetrailing edge 10 at of the transmitted pulse 10 a. Secondly, the DCcomponent of the base line error and the offset errors of the amplifiersof the ultrasonic transmit-receive unit 10 pass through the signal pathand are amplified. This can cause various dynamic range and saturationproblems described further on.

Conventionally, flaw detectors have provisions that allow the user tooperate the instrument either with filters or through direct coupling inorder to select the optimal setting for the flaw measurement scenario.

Reference is now made to FIG. 4 to describe detection of flaws near therear surface of the object 14. In some cases, a flaw 14 d may be veryclose to far surface 14 a of target object 14 which will place the flawecho 11 b in close proximity to back wall echo 11 a. In order to performa proper inspection (in accordance with many formal inspectionprocedures), the peak of back wall echo 11 a must remain visible on thewaveform display 18 at all times. The reasons for this are: 1) smallflaws 2 d in target object 14 caused by porosity or materialcontaminants may generate flaw echoes that are not large enough to beseen on the waveform display 18, but may reduce the amplitude of theecho traveling to back wall 14 a, thereby causing the amplitude of flawecho 11 b and back wall echo 11 a to be reduced, and 2) probe 12 may beimproperly coupled to the surface 14 c of target object 14intermittently, thereby reducing the amplitude of back wall echo 11 a.These two conditions may cause the echo of flaw 14 d to not be visibleon the waveform display 18. However, the reduction in back wall echo 11a will indicate a problem with the target object 14 material or thecoupling of probe 12. If the peak of back wall echo 11 a was allowed togo beyond the top visible portion of the waveform display 18, areduction in peak amplitude may not be visible on waveform display 18.The person performing the inspection sets up the back wall echo 11 adetection parameters by adjusting back wall echo gate 6 d (see FIG. 4)to set the region on the horizontal time axis where the back wall echo11 a is permissible. A threshold on the vertical amplitude axis is alsoset for the minimum acceptable echo amplitude. Typically, an alarm willoccur when back wall echo 11 a falls out of these parameters.

This measurement method produces certain problems.

The difference in echo amplitude between the flaw echo 11 b and backwall echo 11 a can be huge (as much as several orders of magnitude). Butseveral methods described below (a, b, c and d) can be used to ensurethat flaw echo 11 b and the peak of back wall echo 11 a both remainvisible on waveform display 18:

(a) Connect probe 12 to two parallel receiver and A/D converter channels(A and B). The gain of channel A is adjusted by the person performingthe inspection to optimize the amplitude of the echo of flaw 14 d tomake it is clearly visible on waveform display 18. The gain of channel Bis adjusted to ensure that the peak of back wall 11 a echo remainsvisible on waveform display 18 for the reason previously described.

The digital outputs of the channels A and B A/D converters are combinedin such a way that the entire horizontal time scale of waveform display18 shows all of the output of the channel A except for the region ofback wall echo gate 6 d. The leftmost side of back wall echo gate 6 dindicates the point in time when the switch over from channel A tochannel B would occur.

Unfortunately, the two channel method has disadvantages. Typically, aninspection is performed by moving probe 12 along the surface of targetobject 14 in a scanning motion because the presence or location of aflaw inside of the target object is not known until it is detected. Ifthe target object does not have a constant thickness between frontsurface 14 c and back surface 14 a in the scanning area, the back wallecho gate 6 d will need to be adjusted wide enough to include thisvariation in thickness in order not to miss the detection of back wallecho 11 a.

Consequently, near back wall flaw echo 11 b will not be detected if itis very close to back surface 14 a because back wall flaw echo 11 b willoccur within the region of back wall echo gate 6 d. This causes anundesirable effect on near surface resolution by far surface 14 a.Further, the amount of receiver hardware is approximately twice as muchas is required for a single channel solution.

(b) The two successive pulse-receive measurement cycles method issimilar in concept to the two parallel receivers and A/D converterchannels method except only one channel is required. The description insection (a) above applies to the two successive pulse-receivemeasurement cycles method. Also, instead of processing the flaw echo 11b and back wall echo 11 a in two parallel channels set to differentgains, the echoes are processed in the same channel, one pulse-receivecycle after the other, but with a different gain for each cycle.

A disadvantage that is unique to the successive pulse-receivemeasurement cycles method is that flaw echo 11 b is separated in timefrom back wall echo 11 a by an additional pulse interval To (shown inFIG. 2). Therefore, measurement errors are more likely to occur whenprobe 12 is moved because its location may change between the time thatflaw echo 11 b and back wall echo 11 a are measured.

(c) Time varied gain (TVG) is a single channel solution wherein the gainof the amplifiers of the ultrasonic transmit-receive unit 10 isdynamically changed to optimize the amplitude of flaw echo 11 b and backwall echo 11 a (for the reason already described).

The TVG method has the same disadvantage for near surface resolution byfar surface 14 a as the two parallel receiver and A/D converter channelsmethod does.

But there are other disadvantages associated with the TVG method. Thus,FIG. 5 shows an ideal TVG curve 6 e that changes instantaneously fromgain 6 f to gain 6 h, thereby introducing no additional near surfaceresolution error from the analog TVG amplifier. The error associatedwith measuring a flaw near the back wall of a target object withnon-constant thickness, as described in method a above, would stillremain.

Unfortunately, it is impossible for analog TVG amplifiers to achieve theideal curve 6 e (especially, the instantaneous slope 6 g). Analog TVGamplifiers and the external signals that control them have responsetimes that limit gain rate of change 6 g, thereby causing an undesirableeffect on the near surface resolution by far surface 14 a. The nearsurface resolution degrades because flaw 14 d must be farther away fromrear surface 14 c of target object 14 in order to provide time interval6 m for the gain to change. Stated in terms of the echoes of interest,flaw echo 11 b must occur before the start of time interval 6 m, andback wall echo 11 a must not occur before the end of time interval 6 m.

The other problem associated with the TVG method is caused by thevarious sources of DC offset errors in the receiver section ofultrasonic transmit-receive unit 10. The sources include the input DCoffset errors of the amplifier IC's and the DC component of the baseline error.

The DC offset errors present in certain existing flaw detectors of thepresent assignee are compensated for at each gain setting every time thegain is adjusted from one level to the next. The DC offset errors arecompensated for in this way to take into account the effects oftemperature, and long term stability, drift on the DC offset errors,etc. The compensation method uses several D/A converters along thereceiver signal path to inject a DC null signal that will ensure thatthe base line remains on the center of the A/D converter's full scalerange and in an optimal location on waveform display 18. Every time theinstrument is turned on, or the gain setting is changed, an algorithmruns in a microprocessor that takes a base line error reading,calculates the DC error correction value required, and sets the DACs tothis value.

It is not practical to perform the DC offset compensation methoddescribed above for every gain setting at the speed that the TVG isrequired to operate at. Instead, the DC offset correction is set for themidpoint gain, thereby splitting the error between the end points. Forexample, if the TVG range is set to operate between 20 and 60 dB, the DCoffset correction is set to compensate for the error at 40 dB. Theproblem with this technique is that it introduces errors to echoamplitudes that are undesirable for an accurate flaw detection andsizing.

(d) Logarithmic amplifiers are used to cover the huge dynamic rangerequired and the echoes are shown on the waveform display 18 on alogarithmic scale. The logarithmic scale provides a very high dynamicrange thereby allowing both a low amplitude flaw echo and the peak ofthe much higher amplitude back wall echo to be visible on a waveformdisplay.

Unfortunately, certain undesirable consequences occur when using thelogarithmic method. Thus, for a given back wall echo amplitude andamplitude variation, the vertical variation of the peak of the echowaveform is much less noticeable on the waveform display than for areceiver that uses linear amplifiers. This would make it more difficultto detect a flaw by observing the peak amplitude variation of the backwall echo, as described earlier.

Further, the output of the logarithmic amplifier can only provide arectified waveform. Therefore, the location of the negative echo lobecannot be identified because it is either removed by half-waverectification, or converted into a positive lobe by full waverectification. The precise location of both the positive and negativeecho lobes is very important for measuring the thickness of targetobject 14 accurately because one lobe may be more visible than theother. The polarity of the echo lobes is also required to determine whenecho phase inversion occurs. Phase inversion of an ultrasonic echooccurs when a sound wave passes from a material of low acousticimpedance to a material of high acoustic impedance.

Furthermore, all filters must be located prior to the logarithmicamplifier section because the filters require linear signals to operatecorrectly (a logarithmic amplifier is a non-linear device). The receiverwill have a much higher susceptibility to noise if the filter circuitsare located prior to the high gain logarithmic amplifier section becausethe PCB traces required to connect the filter components together aresusceptible to electromagnetic noise, and the internal noise generatedby an filter amplifiers will be maximally amplified. These problems withlogarithmic amplifiers are ameliorated in the present invention becausethe full dynamic range of the sampled data is provided on every sampleclock cycle, thereby allowing it to be rendered as linear scale orlogarithmic scale. Accordingly, the present invention enables anoperator to instruct the system, eg the FPGA described furtheron, toselect and develop for display on display 18 either a linear orlogarithmic system output or to store such outputs for later analysis.

This present invention aims to ameliorate or avoid the drawbacks of theprior art and, in effect, is essentially equivalent to a 100 MHz 24 bitA/D converter that works with a large input voltage, free of the DCoffsets, base line errors and other drawbacks of the prior art. It isimportant to note that although the invention was implemented withperformance essentially equivalent to a 100 MHz 24 bit A/D converter, asdescribed above, it may also be implemented with a sampling frequencyand resolution other than 100 MHz and 24 bits respectively. It utilizesthree (or more) A/D converters operating in a corresponding number ofchannels. The instant inventors recognize that the eventual developmentof multifunction operating A/D converters will permit the use of a lowernumber of A/D converters.

A more detailed version of a prior art circuit which has been utilizedto implement an ultrasonic inspection system is illustrated in blockdiagram form in FIG. 6. This intensely analog circuit utilizes thesignal from the transducer 12 to feed it through a switch 24 as oneselectable input to a series of parallelly providedamplifiers/attenuators 28, 30, 32, 34 and 36, which have respectivegains of 14 dB, 0 dB, −8 dB, −14 dB and −20 dB, respectively. The switch24 also receives the input of a gain calibrator 20 and provides itssignal directly to attenuators 32, 34 and 36, and via switch 26 to theamplifiers 28 and 30.

Variable gain amplifiers (VGA) 40, 42 and 44 respectively receive theirinputs from the amplifiers 28 and 30 and from the switch 29, whichprovides an output 31 that constitutes the selected one of the outputsof attenuators 32, 34 and 36. The outputs of the VGAs are provided to aswitch 46 which also receives as one of its inputs, a signal from gaincalibrator 22 and selectively providing these signals over a bus line 48to a series of high pass filters 50, 52, 54, 56, 58, 60, 62 and 64,whose outputs are switched through a switching network 66 to low passfilters 70, 72, 74, 76, 78, 80, 82 and 84. Thus, the signals from theVGAs 40, 42 and 44 or from the gain calibrator 22 can be fed bycontrolling the selection o a desired signal through the switches 66 and67 to provide it to a further, downstream VGA 86, whose output isfurther provided through a switch 92 to an amplifier 90.

The output of amplifier 90 or the output of a gain calibrator 94 arethen finally fed to the 100 MHz 10 bit analog to digital (A/D) converter100.

A field programmable gate array (FPGA) 106 incorporates a real timesample data control and storage circuit 102, and a measurement gaindetection and compression circuit 104 to provide an output to thedigital signal processor and control, which also controls the settingsof the FPGA 106 to obtain the appropriately processed output of theanalog to digital converter 100, provides time varied gain control, andproduces a signal that can be displayed on the display 18.

In view of the introductory discussion, it is readily apparent that thetasks of calibrating the various analog circuits to preventinconsistencies and variations attributable to different frequencyresponses of the numerous high pass and low pass filters, and avoidingthe DC offsets and drifts and a temperature effects of the analogdevices present enormous challenges to both designers and users of theprior art circuits.

A cursory comparison of the block diagram of the present inventionpresented in FIG. 7 illustrates the far scarcer usage of the problemprone analog circuits in the instant invention, which utilize triple A/Dchannels that avoid many of the drawbacks and complexities of the priorart.

In the block diagram of FIG. 7, when switch 114 a is closed thetransducer 12 has its output 13 a provided directly to only twopreamplifiers 110 and 112, the latter amplifier feeding a thirdamplifier 122. The signals of these amplifiers are processed,respectively, in frequency response trim and filter blocks 116, 118 and120 and subsequently provided along the three channels A, B, C todifferential amplifier drivers 126, 128 and 130. The analog signalsalong the three channels are then provided directly to A/D converters132, 134 and 136, respectively, whose digital outputs in turn are thensupplied to the field programmable gate array 140, which incorporatescontrol and storage block 142, time varied gain, and measurement gatedetection and composite A-scan compression circuit 152. This FPGA 140works in conjunction with the DSP 160, which provides its signal to thedisplay 18.

The implementation in FIG. 7 (the functionality and features of whichare discussed in greater detail below in relation to the description ofFIGS. 8 a and 8 b) dispenses with most of the analog circuits and thedrawbacks of the prior art, including the intensive use of analog highpass and low pass filters, additional amplifiers and calibrators, andvarious VGA circuits, all of which are rendered unnecessary inaccordance with the circuits of FIGS. 7, 8 a and 8 b.

Thus, as further shown in FIGS. 8 a and 8 b, the present invention is anapparatus and method for extending the dynamic range of an A/D convertercircuit used in a flaw detector, thickness or corrosion measurementinstrument that eliminates the need for a Variable Gain Amplifier (VGA)circuit and its associated complexity and performance limitations. Theapparatus and method of the invention utilize three A/D converters thatsample on different channels three differently scaled versions of thesame input signal. The sample times of each channel are adjusted tocompensate for the propagation delays of each preamplifier channel tominimize signal skew errors between the sample data output of each A/Dconverter. The scaling is such that the maximum gain channel (C) has aresolution that is 32 times higher than the mid-gain channel (B), and1024 times higher than the minimum gain channel (A). The higherresolution channels are monitored for data overflow, and the channelthat has the highest resolution data without overflow is selected as theoutput. The selected outputs are merged to produce a seamless stream ofoutput data. The resultant output is a stream of data in which thequantization step-size is larger for large signals and 32 or 1024 timessmaller for small signals. The level of dynamic range thus provided bythe present invention eliminates the implementation of the traditionalVGA to control the level of the analog input signal to keep the peakvoltage level of the analog input signal at or near the full-scale valueof the input for the A/D converter.

When sampled with the circuit of FIGS. 8 a and 8 b, the input signalfrom the transducer 12 is split into two channels 19 a and 19 b, withrespective buffers dedicated for each respective channel. Thus,respective preamplifiers 110 and 112 amplify the input signal 13 a onrespective channels with a gain of 0.1 (−20 dB) and a gain of 3.2 (10.1db), respectively. The output of preamplifier 112 is connected to theinput of preamplifier 122 to create the third channel which has a gainof 102.4 (40.2 dB). Each channel is sampled by one of threesubstantially identical A/D converters 132, 134, 136. The three channelsA, B, C are sampled with time delays between them to compensate forinput signal timing skew errors caused by the propagation delays of allof the amplifiers in the analog signal path. The time delays arecontrolled by the rising edges of the clocks CLKA, CLKB, CLKC that drivethe A/D converters, which clocks are adjusted by a calibrationalgorithm.

In embodiments that have been reduced to practice, sample timingadjustment is separated into two parts.

A) Course adjustment: using one FIFO and control circuit for each A/Dchannel, data is delayed by a selectable integer number of clock cycles.

B) Fine adjustment: There are four Phase locked loops (PLL) running 0,90, 180 and 270 phase angles relative to the clock. By independentlyselecting a PLL output for each A/D, the clock timing of each A/D can beadjusted in steps of ¼ of a clock cycle.

If the converted data of the maximum gain channel (C) is valid, then itsresult is passed through unmodified as output 132 OUT of the threechannel A/D converter circuit (FIG. 8 b). If the converted data of themaximum gain channel (C) has overflowed, then its result is discarded,and the result of the converted data of the mid-gain channel (B), if ithas not overflowed, is passed through, scaled to correct for the gain ofpreamplifier 112 and used as output 134 OUT. If the converted data ofthe mid-gain channel (B) has overflowed, then its result is discarded aswell, and the result of the converted data of the minimum gain channelis scaled to correct for the signal path gain. This scaled gain iscalculated as:

Gains of: preamplifier 112+preamplifier 122−preamplifier 110, which isthen used as output 136 OUT.

In the embodiment illustrated in FIGS. 8 a and 8 b, the three channelA/D converter circuit of the invention is capable of: eliminating signaloffset errors in all three separate channels; scaling the input signalby using three independent preamplifier channels each set to a differentgain; converting the analog signal input to each of three separatechannels to a digital signal at respective sample times that areadjustable to compensate for input signal timing skew errors; detectinga channel overflow condition at least in channels having higher gains;and merging the A/D converter outputs of the three channels in realtime.

As noted above, the analog input signal 13 a from the transducer 12 isdirected to two signal clamping amplifier channels, characterized by twoclamping elements 111 a and 111 b, where the second amplifier 112 of thetwo amplifier channels has a gain that is higher than the gain of thefirst channel 110 by a predetermined factor. The output of the channel Bamplifier 112 is connected both to the downstream filter 118 and also toamplifier 122 with a gain of 32 to create channel C. For example,channel A has a gain of 0.1, while channel B has a gain of 3.2, andchannel C has a gain of 102.4. Thus, compared to each other, channel Aand B differ by a gain factor of 32, channel C and B differ by a gainfactor of 32, and channel A and C differ by a gain factor of 1024.

Clamping voltage thresholds for amplifiers 110 and 112 are set to levelssuch that the resulting output slightly exceeds the valid input range ofrespective channel A, B and C A/D converters 132, 134 and 136. The clampcircuits 111 a, 111 b, and 113 also limit the input voltage to the gainchannel amplifiers to prevent them from entering into saturation.

The prevention of amplifier saturation is important because once insaturation, it takes a considerable amount of time for an amplifier toreturn to its linear region of operation. By preventing the amplifiersin the gain channels from becoming saturated, the length of time thehigher gain A/D converters are in the overflow condition is minimized,thereby allowing the higher resolution output data to be used sooner.The clamp circuit in preamp 112 also serves to maintain a constant inputimpedance for input signal 19 a, regardless of the input signal level upto a signal level higher than the maximum input to channel A preamp 110.The input signal would become distorted if a constant input impedancewas not maintained.

The present inventor recognizes that amplifier 122 does not requireclamp 113 to maintain a constant input impedance for transducer 12 overits signal amplitude operating range because amplifier 122 is isolatedfrom transducer 12 by means of amplifier 112. Because of this, anotheramplifier circuit configuration can be used for amplifier 122 if neededto provide other benefits such as lower power or less circuitcomplexity.

In an embodiment that has been reduced to practice, the channel Camplifier 122 is allowed to saturate and fast recovery OpAmps are used.Preferably, clamping may be added to generate less noise.

The output of each gain channel amplifier 110, 112, 122 is connected toFrequency Response Trim and Filter circuits 116, 118, 120, respectively.Frequency response adjustment control signals 116 a, 118 a, 120 a, arerespectively used to make the frequency response of channels A, B and Cmatch as closely as possible. This is required to make sure that allsignal frequencies of interest have as close to the same gain aspossible. A calibration algorithm is used to adjust the frequencyresponse as described above. This frequency trimming method may be usedfor two or more analog to digital converter channels.

The anti-aliasing filter function for channels A, B and C is distributedwithin Frequency Response Trim & Filters 116, 118 and 120 andDifferential Amplifiers 126, 128 and 130 respectively.

The DC offsets inherent in the amplifiers of each channel arecompensated for by injecting DC signals 112 a, 122 a, 126 a and 128 a tocounterbalance the DC offset errors present throughout the analog signalpath. A calibration algorithm is used to perform this compensation. Itshould be noted that this DC offset compensation method has thefollowing two limitations:

-   -   1) At very fast pulser/receiver repetition rates (To of FIG. 2),        there is not sufficient time available between To cycles to        perform the DC offset correction process required to compensate        for DC offset drift over time. This limits the DC offset        calibration to only occur when the instrument is not measuring.    -   2) At very high gain settings, the small DC offset error that        remains after counterbalancing will produce a significant offset        in the stored sample data, and subsequently in the waveform that        appears on the display.

To further ameliorate the effects of DC offset errors present throughoutthe analog signal path, including the effects described in items 1 and 2above, the present embodiment includes the purely digital DC offsetcompensation method, a block diagram of which is shown FIG. 8 c.

With further reference to FIG. 8 c, the output of A/D converter 136 isprovided to Base Line Capture block 146 during the interval 10 c shownin FIG. 3. Sample points from interval 10 c are used to monitor the baseline because they are in a relatively ‘quiet’ region in time—i.e. aregion that occurs before the pulser fires and after ultrasonic responsesignals of substantial amplitude will be present. In the presentembodiment, Base Line Capture block 146 uses 256 signed integer samplepoints and calculates the average; however, a different number of samplepoints may be used. When multiplexer 147 is enabled by control signal149 to allow the signed integer output of Base Line Capture block 146 topass through to Base Line Corrector block 148, signal 147 a issubtracted from signed integer signal 145 a to remove the base lineerror. Register 150 is intended to allow alternate base linecompensation value to be used that may have been produced by a softwarealgorithm or hardware device not shown.

The A/D converters 132, 134 and 136 of the three channels are 14 bit,high speed converters for which sample timing is provided by the sampleclocks CLKA, CLKB, CLKC derived from a 100 MHz oscillator block 131using respective delay control elements contained within a FPGA circuit.The delay control elements enable adjusting the placement of the risingedges of the sample clock of one channel in time with respect to a clockcircuit portion of another channel so that the sample times of eachchannel are adjusted to compensate for the propagation delays of eachpreamplifier channel and any other source of timing skew revealed byexamination of the A/D converter output data. A calibration algorithm isused to perform this compensation.

As previously noted, in embodiments that have been reduced to practice,sample timing adjustment is separated into two parts.

-   -   1) Course adjustment: using one FIFO and control circuit for        each A/D channel, data is delayed by a selectable integer number        of clock cycles.    -   2) Fine adjustment: There are four Phase locked loops (PLL)        running 0, 90, 180 and 270 phase angles relative to the clock.        By independently selecting a PLL output for each A/D, the clock        timing of each A/D can be adjusted in steps of ¼ of a clock        cycle.

The present inventor contemplates an alternate method of adjusting thesample data timing by use of a fine analog adjustment in conjunctionwith the course digital adjustment described above. An adjustable signaldelay element would be used to adjust the timing of the analog signalsinstead of the digital clock timing adjustment method described above.This analog signal delay could be accomplished by using any one of thefollowing methods.

-   -   1) A delay line with taps, a tap is selected by a switch to        adjust the delay.    -   2) Delay filter elements switched in or out of the signal path        as needed.    -   3) An adjustable delay constructed using a variable element such        as an all pass delay filter using a voltage controlled        component. The delay could be controlled by a DAC to provide        very fine control. The present inventor perceives this method to        provide the best adjustment resolution.

A method is also provided to calibrate the system gain by adjusting thefull scale range of A/D converters 132, 134 and 136. This isaccomplished by adjusting the reference voltage (not shown) of therespective A/D converters using D/A converters (not shown). Acalibration algorithm is used to perform this function.

The digital outputs of the A/D converters 132, 134 and 136 are connectedto digital multiplexing circuit 135. The overflow signals for the twohigher gain A/D converters 134 and 136 are connected to Channel SelectLogic circuit 137. Channel Select Logic circuit 137 also extends thetime duration of overflow signals from A/D converters 134 and 136 inorder to provide time for all of the amplifier circuitry prior to theinput of A/D converters 134 and 136 to come out of saturation. Thiscircuit 137 selects the output data bus from the highest gain channelA/D converter that has not overflowed. If all three A/D converterchannels are overflowed, the output data bus of the lowest gain channelA/D converter is selected because it will be the first channel to comeout of the overflow condition.

Channel Select Logic circuit 137 and the overflow signal from A/Dconverter 132 are connected to an Exponent Generator circuit 139. Thiscircuit 139 calculates the exponent to go with the selected A/Dconverter data in RAM 141. A floating point conversion circuit 143effectively adds bits of precision to the A/D conversion for smallsignals, while maintaining the range capacity for large signals. Thefloating point converter 143 also reduces the number of bits the sampledata RAM requires. The sample data RAM has 18 bits of which 14 bits areused for the mantissa, and 4 bits are used for the exponent. When asampled value is stored, the selected A/D converter value is stored inthe mantissa and an exponent value of 0, 5, or 10 is stored in theexponent to indicate the scale of the data. The exponent may also be setto 15 to indicate that all channels are in the overflow condition.Furthermore, when the data is read from the sample RAM 141, the exponentis used to position the data in the mantissa to construct a 24 bitinteger output of the Floating Point to Integer Converter 143. This isthe final output 145 of the present invention. This output can berepresented by the following formula:Output 145=2^(exponent)×mantissa=24 bit integer

Although the present invention has been described in relation to anembodiment which utilizes three signal processing channels, eachincorporating its respective analog to digital converter, the instantinventor also contemplates the use of a lesser number of analog todigital converters or even a single analog to digital converter. Thus,for example, if an analog to digital converter operating at 200 MHzbecomes available, two of the channels may be handled by a single analogto digital converter that produces two successive rapid samples of thesame signal point. To do so, a first sample of a signal may be taken,while an amplified version of the same signal is delayed (by an analogdelay time) for a time delay approximately equal to the clock period ofthe 200 MHz analog to digital converter. Then the Delayed amplifiedsignal is sampled by the same A/D converter. Also, analog comparatorsmay be utilized to compare the signal magnitudes at the outputs of thepreamplifiers to determine their magnitude ranges and to controlchanneling the signal to that one of the analog to digital converterswhich will not overflow in response to that signal magnitude.

Further, while three channels have been utilized, it is within theconcept of the invention to utilize four or more channels for thepurpose of increasing the overall dynamic range of the testing systemand/or for a purpose of using a given analog to digital converter as atemporary substitute for any one of the analog to digital converterswhich has temporarily overflowed by having been saturated.

Elaborating on the aforementioned extensions of the present invention,one implementation may be in the form of a two channel system using apair of 16 bit ultra fast analog to digital converters, the clock speedsof which are sufficient for applications of the present invention. Notefurther, that the full dynamic range is not always required in everyapplication, as specific users may require less than a full dynamicrange and so may be able to use only one of the multiple analog todigital converter channels. In a two channel system, with one of thechannels switched between low gain and high gain, it would be possibleto provide a good portion of the benefits of a three channel system,utilizing only two channels.

Relative to the aforementioned problem of detecting flaw echoes that arevery close to the back wall of the target object, the present inventorrecognizes that the problem can be solved if both channels are storedand the channel change is carried out in the post processing. This wouldbe a ‘tracking back wall attenuator’ solution. Also a dual or splitdisplay window could be used, one to show flaws and the other the backwall. This would remove the need to track the back wall and adjust thedisplay. A small section of the received signal would be displayedtwice—once at high gain in the flaw section and then again at low gainin the back wall section. This method can only support a flaw alarm gatethat detects flaws that are very close to the back wall if the gateposition is calculated in the post processing.

Relative to the aforementioned concept of adjusting the frequencyresponse of the channels individually to make the assembled data streamfit together without steps or jumps, it should be noted that this can bedone using a factory adjustment or a run time adjustment. Note further,that in a three channel system, it may be sufficient to provide thefrequency response trims on only two of the channels.

The invention can also be implemented by extending the duration of theover range indication signal to prevent output data of an analog todigital converter from being selected prior to the time when its signalchannel has fully recovered from the saturation condition. This can takeone or more of the following forms.

1. In a current embodiment, time is added to the end of the over rangeindicator bit from the analog to digital converter. This feature isimplemented within the channel select logic 137, shown in FIG. 8 b. Itmay consist of an OR gate that receives the overflow signal as one inputand a shifted version thereof as another input.

2. A digital comparator is used on the channel with the next lower gainto detect when the analog to digital converter is out of severesaturation, even if the analog to digital converter still indicates overrange. Adding delay to this “severe saturation” detector is comparableto providing a delay on the over range indicator.

3. The data out of the analog to digital converter is compared to thevalue of the next lower gain channel to validate the data. The valuemust be within a prescribed range of the value from the next channel.

4. An analog to digital converter is used that is slow to indicate thatit has come out of over range.

It should be noted further that an analog to digital converter maysaturate at an input voltage that is higher than over range voltage.This is why providing a delay coming out of saturation is advantageous,whereas a delay coming out of over range is not needed. In the voltagerange between over range and saturation, the analog to digital convertermay function normally and not need recovery time. In an embodiment thathas been reduced to practice, the analog to digital over range indicatorhas been used as a saturation indicator and will sometimes introduce adelay that is not needed. This unwanted delay is rare and not of anytechnical significance.

The present inventor also contemplates the use of digital to analogconverters to trim the analog to digital converter reference voltagesfor the effect of trimming the gain. This method is used to extend therange of the user gain control and is different from channel matching.

The present inventor also contemplates the use of a preamp for the midand high gain channels that does not distort the source signal. Thisapproach is preferable to building or utilizing an amplifier with atleast a 20 volt peak output range of very low noise performance. Thedescribed approach is also preferable to a hybrid design, where theinput uses attenuator steps, but this approach does not have as muchdynamic range. Nonetheless, for a low cost market segment, a hybriddesign may be preferred.

In the preceding description, reference was made to various technicalconcerns relating to circuit devices becoming saturated or analog todigital converters indicating over range. Following an initialdiscussion of the underlying problem, several alternate solutionsrepresenting further embodiments of the invention are provided.

Under normal operating conditions, the channel gains for the circuitsidentified below in brackets are applicable.Channel A Gain*32≈Channel B Gain  [FIG. 7]Channel B Gain*32≈Channel C Gain  [FIG. 7]

When a channel is driven into saturation it will be indicated by theoverflow output signal of the channel's analog to digital converter,thereby enabling Channel Select Logic 137 to select the best channel toreceive the signal. As previously described, the best channel is the onewith the highest gain and not in the overflow state. The lowest to thehighest gain is Channel A, Channel B and Channel C, respectively. SeeFIGS. 8 b, 8 c, 8 d, and 8 e.

Any, or all, of the above conditions may not be true for very fast slewrate signals, such as the leading edge of the pulser pulse, because theedge is so fast that the amplifiers of all three channels are driveninto saturation at substantially the same time.

Due to the slew rate limitations of amplifiers and filters, the analogto digital converters do not saturate immediately, and all threechannels move toward saturation at substantially the same rate. Ifsamples are taken from the A/D, while their outputs are slewing to theirfinal values, erroneous readings will be noted. For example, when allthree channels are at about ½ full scale (corresponding to an A/D outputvalue (in HEX) of 2FFFC), they will not correspond to the correct inputamplitude. The channel readings, none of which indicate an overflow,would be as follows:Channel A=2FFF, indicating −5V at the input.Channel B=2FFF, indicating −0.15V at the input.Channel C=2FFF, indicating −0.005V at the input.

Therefore, the embodiment of FIGS. 8 b and 8 c will select Channel Cbecause it is the channel with the highest gain and not (yet) in theoverflow state. The channel readings above indicate that that Channel Ais −5V, or less; therefore, the −0.005V signal (assumed to be at theinput of Channel C) would be shown on the display, which would beincorrect.

As shown in FIGS. 8 d and 8 e, an alternate embodiment does not requirethe use of an overflow output signal from any of analog to digitalconverters 132, 134 and 136. Instead, magnitude comparators 801, 802 and803, respectively, are used to indicate when the digital output data ofeach analog to digital converter matches a predetermined number.Magnitude comparators 801, 802 and 803 provide an output signal toChannel Select Logic 137 when the predetermined number is matched foreach. Magnitude comparator 801 also provides its output signal toExponent Generator 139. It should be noted that the performance of thepresent embodiment can also be achieved by using only magnitudecomparators 801 and 802 for channels A and B, respectively.

Due to the fact that the digital output signal of a channel's analog todigital converter can be correlated to the level of a signal at anypoint along the input signal path, the primary advantage of the‘magnitude comparator’ method is that it can be used to detect anysignal level of interest that falls within the full scale of the analogto digital converter and is within its measurement resolutioncapability. A saturation condition of an amplifier within the inputsignal path is one example of a signal level of interest.

Referring to FIG. 10, when processing a very fast signal edge (i.e. fastslew rate) the following logic is true. It should be understood that thevalues shown below are 14 bit signed integers.

-   -   a) IF [Channel A>=100] or [Channel A<=3EFF], then the Channel B        and Channel C amplifiers are probably over driven.    -   b) IF [Channel B>=100] or [Channel B<=3EFF] Then the Channel C        amplifier is probably over driven.

Using the logic of a) and b) above, the problem of erroneous channelselection can be prevented by incorporating into Channel Select logic137 the following rules in the order of priority show below:

-   -   a) IF [Channel A>=100] or [Channel A<=3EFF], then use data from        Channel A—i.e. Channel A has priority over Channel B    -   b) IF [Channel B>=100] or [Channel B<=3EFF], then use data from        Channel B—i.e. Channel B has priority over Channel A    -   c) IF [Channel A<100 and >3EFF] and [Channel B<100 and >3EFF],        then use data from Channel C—i.e. Channel C has priority over        Channel A and B

It should be noted that the hexadecimal values used above and in FIG. 10were chosen as examples and are not necessarily the values used in anactual embodiment.

FIG. 8 d further illustrates in dashed lines a Channel Blender 135′ usedas an alternative for MUX 135. Channel Blender 135′ serves for blendingthe output of two of the three A/D Converters that have the highestgain, but are not in saturation, in order to minimize the effects ofunmatched signals between channels.

FIG. 11 is approximately equivalent to the circuits and signalscontained within Channel Blender 135′; however, it only shows theportion for channel A and B, and more output circuits would need to beadded to be compatible with the input required for RAM 141.

As used herein, “blending” refers to combining or associating, so thatthe separate constituents or the line of demarcation is not easilydistinguished. Thus, Channel Blender 135′ is a device that takes outputvalues from two adjacent A/D converter channels and calculates acompromise value as its output. A ratio control is needed to control theproportions of the two inputs used.

FIG. 11 shows the details of the ratio control circuits.

In this example, the value of the ratio control is limited to the rangeof 0 to 1.(Input A)*Ratio+(Input B)*(1−Ratio)=Output

For reasons of circuit simplicity, the ratio control may be furtherlimited to a small set of discrete values that may not include 0and/or 1. The numbers 0 and 1 produce an output that is identical to oneor the other input; some other circuit may handle this condition.

A very simple blender assembled from two adders and three multiplexerscould support the following ratio values: 0, 0.25, 0.5, 0.75 and 1. Thiswould divide the channel select anomaly into four separate anomalies,each of one-fourth the magnitude.

Thus, depending on the amplitude of input signal 19 a of FIG. 7, ChannelSelect Logic 137 of FIG. 8 d selects the active channel. When thissystem is used for an application that produces an input signalamplitude very close to a threshold that causes the system to switchchannels, it may be observed that, as the system changes channels, asmall jump, or glitch, might appear in the output because the gain,frequency response and/or phase of the two channels are not preciselymatched. This may manifest itself as an unexpected rise or fall in theoutput signal amplitude.

Referring to FIG. 8 d, the low gain channel is channel A and the highgain channel is channel B. Depending on the outputs of MagnitudeComparators 1102 and 1108 (FIG. 11) contained within Channel Blender135′ (FIG. 8 d), Blender 1111 (FIG. 11) measures how close to saturationchannel B is. As input signal 19 a (FIG. 7) is increased, channel Bapproaches saturation and the preset value of Magnitude Comparator 1108.When the latter is reached, a blending function is invoked withinChannel Blender 135′ that mixes the data from A/D converter 134 withdata from A/D converter 132. The blend function is variable or has stepsof weighting of the two data sources from the respective A/D converters.As channel B approaches saturation the blend-weighting ratio is changedso as to apply more weight to channel A and less to channel B. As anexample: starting at a low input signal 19 a (FIG. 7) amplitude, theblend ration would be at 100% of channel B and 0% of channel A; aschannel B gets closer to saturation, the blend would change to 50% ofchannel B and 50% of channel A; when channel B is saturated, the blendis 100% of channel A and 0% of channel B. The blend ratio can beextracted from channel A or B or a combination there of. The blend ratiomay change in a few steps or adjust smoothly in proportion to thechannel signal amplitudes.

The use of Channel Blender 135′ makes the input signal 19 a (FIG. 7)voltage threshold that separates channel A operation from channel Boperation less likely to be observed by the operator. This blendfunction can be used for all channel transition points. This method canbe used in combination with any of the other methods of controllingchannel selection.

FIG. 8 h provides another solution which adds delay elements 808, 810,811, 813, 814 and 816 to the output sample data and overflow (OF)signals of analog to digital converters 132, 134 and 136 in order toprovide the overflow signals an additional clock cycle to respond beforethe output sample data is provided to MUX 135. Although not shown, morethan one additional clock cycle may be used. The delays prevent ChannelSelect Logic 137 from selecting a channel until the overflow signalshave had sufficient time to respond, thereby preventing the problemcaused by a fast slew rate input signals that was described earlier.

In each channel, the overflow and delayed overflow signal are providedto an OR gate in order for the overflow signal to:

-   -   a) Turn on without delay so that it will occur prior to the time        when the analog to digital converter output sample data is        provided to MUX 135, and    -   b) Turn off with delay in order to be synchronized with the        delayed sample data that is provided to the input of MUX 135        upon returning from the overflow state.

It should be noted that delays of other than one clock cycle may also beused for this alternate embodiment.

This method is implemented by inserting a data delay of one sample clockcycle between each digital signal output of analog to digital converters132, 134 and 136, and the input to MUX 135. A data delay of 1 sampleclock is also inserted between the overflow signal of each analog todigital converter and the input to an OR gate. The output of OR gate 809of channel A is provided to the input of Exponent Generator 139. Theoutput of OR gate 812 and 815 for channel B and C, respectively, areprovided to Channel Select Logic 137.

It should be noted that the performance of this alternate embodiment canalso be achieved by using delays in channels B and C only.

The present inventor also contemplates a method to make the gain of eachchannel substantially meet predetermined levels by using only a variablegain mechanism, such as a variable gain amplifier, within in the analogsignal path of each channel. The gain levels would be set topredetermined levels by a calibration procedure. The predeterminedlevels contemplated for the present embodiment are those levels thatensure that the gain scaling between channels A, B and C is as accurateas possible. There are no figures associated with this alternateembodiment.

In the preceding description, reference is made to a base line corrector148, shown in FIG. 8 c. As described below, digital DC offset adjustmentcan be performed at the output of any of the analog to digitalconverters, instead of just at the merged output, as shown in FIG. 8 c.Accordingly, reference is now made to FIGS. 8 e, 8 f and 8 g, noting thefollowing:

-   -   a) Base Line Correction System (BLCS) 804 shown in FIG. 8 f is        the same as what is shown in FIG. 8 e items 146 through 150.    -   b) Base Line Correction Systems (BLCS) 805, 806 and 807 for        channels A, B and C, respectively have the same contents as BLCS        804. BLCS 805, 806 and 807 are redrawn versions of BLCS 804 and        are intended to improve the appearance of FIG. 8 g.    -   c) As shown in FIG. 8 g, BLCS 805, 806 and 807 are inserted        between the digital signal outputs of analog to digital        converters 132, 134 and 136, and the input to MUX 135.

With further reference to FIG. 8 g, the outputs of A/D converters 132,134 and 136 are provided to BLCS 805, 806 and 807 during the interval 10c shown in FIG. 3. Sample points from interval 10 c are used to monitorthe base line because they are in a relatively ‘quiet’ region intime—i.e. a region that occurs before the pulser fires and afterultrasonic response signals of substantial amplitude will be present. Inthe present embodiment, BLCS 805, 806 and 807 each use 256 sample pointsand calculates the average; however, a different number of sample pointsmay be used. The multiplexers within BLCS 805, 806 or 807 can be enabledby their respective control signals (ME) to allow the output of eachBLCS to be provided to Base Line Corrector block input B, as shown inFIG. 8 f. Input B is then subtracted from the output of A/D converters132, 134 and 136 to remove the base line error. The Registers containedin BLCS 805, 806 and 807 are intended to allow alternate base linecompensation value to be used that may have been produced by a softwarealgorithm or hardware device not shown.

The present inventor also contemplates an alternate embodiment, as shownin FIG. 9 and described below, that will achieve the benefits of thepresent invention, especially high dynamic range, by utilizing onesignal path A/D Converter in concert with one or more gain reading A/DConverters and an Automatic Gain Control (AGC) circuit to determine andcontrol the gain of the system. Although not shown in FIG. 9, Inputsignal 10 b of FIG. 1 is connected to Input 200 on FIG. 9.

According to one aspect of the alternate embodiment, a datareconstruction device in Acquisition Logic Block 210 is used to computethe system gain and render the appropriate signal amplitude on thedisplay, or provide it as input to another device. Acquisition LogicBlock 210 would be located within FPGA 140 of FIG. 7 and the circuitryto the left of it would be replaced substantially by the entirety ofFIG. 9. Some of the circuits in FPGA 140 would be modified, or removed,as appropriate per the alternate embodiment.

According to another aspect of the alternate embodiment, the system gainis calculated for every sample point by using the output values ofsignal A/D Converter 209 in conjunction with the output values of gainreading A/D Converters 225 and 226. The sample rate is substantially thesame and synchronous for A/D Converters 209, 225 and 226. The precisionof the system gain calculation is substantially determined by theaccuracy of the gain calibration system, the transfer characteristics ofthe multipliers, and the accuracy of the three aforementioned A/DConverters. The present inventor contemplates that calibration for zeromultiplication (explained later) and DC offset nulling may be requiredfor each channel.

As can further be seen from FIG. 9, the circuit of the alternateembodiment is comprised of four parallel input gain channels 201, 205,207, and 211, the output of each provided to one of four gain controlMultipliers 202, 206, 208, and 212 respectively, the outputs of whichare provided to Adder 203 followed by Amplifier 204, A/D Converter 209,and finally Acquisition Logic 210. AGC circuit 227 receives input fromMonitor signals 213, 214, 215 and 216, and provides output gain controlsignals 217, 218, 219 and 220 to Multipliers 202, 206, 208, and 212respectively. The present inventor recognizes that that number ofchannels may be more or less than four depending on the dynamic rangerequired for the application that this alternate embodiment is appliedto.

The prevention of the undesirable effects of signal saturation that canoccur at different locations along the signal path is a very importantaspect of the alternate embodiment. The signal path starts at Input 200and ends at the input to A/D Converter 209. A saturated signal in thepresent embodiment is considered any signal in the signal path startingat the output of Preamplifiers 201, 205, 207 and 211 that has anamplitude with an absolute value of greater than 1 volt. The followingthree conditions can cause saturated signals to be present in the signalpath.

-   -   1. The absolute value of the amplitude of Input Signal 200 is        greater than 10 V peak.    -   2. The absolute value of the amplitude of Input Signal 200 is        less than or equal to 10 volt peak and of sufficient amplitude        to cause the output of Preamplifier 205, 207 or 211 to be        greater than 1 volt.    -   3. The absolute value of the amplitude of Input Signal 200 is        less than or equal to 10 V peak and the sum of the outputs of        Multipliers 202, 206, 208 and 212 at the output of Adder 203 is        sufficiently high enough to cause signal saturation at the input        of A/D Converter 209.

For condition 1, it is not the object of the alternate embodiment toprevent signal saturation along the signal path because many FlawDetector inspection procedures require that the pulser signal, which hasa peak amplitude absolute value that is much greater than 10 V, toalways be present on the display; therefore, the pulser signal must bepermitted to saturate the signal path.

For condition 2, a means is provided in the alternate embodiment by useof AGC circuit 227 to substantially prevent the saturated output signalsof Preamplifiers 205, 207, and 211 from passing through gain Multipliers206, 208 and 212 by setting Gain control signals 218, 219 and 220 tosubstantially zero. The present inventor recognizes that commerciallyavailable multiplier components do not possess perfect performancecharacteristics. Hence, Multipliers 206, 208 and 212 are not required toprovide the infinite attenuation associated with a theoretical zeromultiplication. Multipliers 206, 208 and 212 are only required toprovide sufficient attenuation to keep the maximum peak amplitude of thesaturated signal below the level that would cause an undesirable effectto the input signal to A/D Converter 209. The maximum permissiblesaturated signal level could be established, for example, from arecognized industry standard for Flaw Detector instruments such as theEN12668-1:2000. It is worth noting that the outputs of Multipliers 206,208 and 212 are summed; therefore calculation of the maximum permissiblesaturated signal level must take this into account.

For condition 3, a means is provided in the alternate embodiment by useof AGC circuit 227 to ensure that the outputs of Multipliers 202, 206,208, and 212 are of sufficiently low amplitude to prevent a signal ofgreater than 1V to occur at the input to A/D Converter 209 after theoutputs have been summed by Adder 203 and amplified by +15 db amplifier204.

According to another aspect of the alternate embodiment, channels A, B,C and D must have substantially equivalent propagation delays andfrequency responses up to, and including, the input of Adder 203 toprevent distortion at the summed output.

According to another aspect of the alternate embodiment, the gain ofeach channel is controlled by multiplier multiplicand signals Gain A,Gain B, Gain C and Gain D which are shown on FIG. 9 as items 217, 218,219 and 220 respectively. Automatic Gain Control circuit 227 monitorsthe output of each gain amplifier by means of Monitor signals 216, 215,214 and 213, and adjusts the gain accordingly. The gain of Multipliers202, 206, 208 and 212 are controlled in a manner to provide a smoothtransition from one multiplier to another thereby preventing abrupt gainchanges that can cause signal distortion or glitches.

According to another aspect of the alternate embodiment, Preamplifier205, 207 or 211, if saturated, is prevented from distorting Input signal200 by use of the clamping circuit previously described for theinvention of FIG. 7. Each clamping circuit prevents distortion of inputsignal 200 by maintaining a constant input impedance for Preamplifiers205, 207 and 211.

According to another aspect of the alternate embodiment, A/D Converters225 and 226 sample the summed Gain signals that are provided to it byAdders 223 and 224 respectively. Gain signals 217 and 219 are eachdivided by ten in order to scale them to match the sensitivity of Gainsignals 218 and 220.

According to another aspect of the alternate embodiment, when the signalamplitude of Input 200 is near zero, the amplitudes of Gain Monitorsignals 213, 214, 215 and 216 will also be near zero, thereby causingAutomatic Gain Control circuit 227 to turn up Gain signals 217, 218, 219and 220 to their maximum gain value of 1 volt. As the signal amplitudeof Input 200 increases, Multipliers with non-zero gain multiplicandschange gradually in order to provide a smooth gain transition betweenchannels prior to reaching a saturation condition. When the amplitude ofInput 200 causes D_Monitor signal 213 to reach a predetermined amplitudejust below saturation, Automatic Gain Control circuit 227 reduces Gain D220 to zero to prevent the saturated signal, when it occurs, frompassing through Channel D multiplier 212 and causing a substantiallysaturated signal. When Gain D is set to zero, Input 200 will passthrough Channel A, B and C until C_Monitor signal 213 reaches apredetermined amplitude just below saturation, thereby causing theAutomatic Gain Control process described above for Channel D to startfor Channel C. As the signal amplitude of Input 200 continues toincrease, this process occurs for Channel B, then Channel A, ultimatelypreventing substantially saturated signals from passing through ChannelsB, C and D.

The response time of AGC circuit 227 establishes the maximum acceptabletime rate of change of the input signal 200 because the gain adjustmentmust occur prior to the moment when input signal 200 reaches theamplitude that will cause an impermissible signal to occur. If thealternate embodiment must work with signals that have a time rate ofchange that is faster than the response time of the AGC circuit 227, adelay circuit is introduced between the output of Preamplifiers 201,205, 207 and 211 and the input to Multipliers 202, 206, 208 and 212.Monitor signals 216, 215, 214 and 213 are connected to the input of eachdelay circuit respectively. The delay circuit provides a time delay thatis greater than the response time of the AGC circuit 227. The relativepropagation delay and frequency response errors between the delaycircuits of each channel must be minimal in order to not cause anunacceptable degree of signal distortion.

The present inventor recognizes that the objectives of the alternateembodiment can be achieved with control parameters and sequences for theAutomatic Gain Control circuit 227 that are implemented in ways otherthan described in the embodiment above. Furthermore, the presentinventor recognizes that these other embodiments may accomplishsubstantially the same end result with respect to gain control.

Throughout the specification and claims, reference is made to “echo”signals. As will be appreciated by people of skill in the art, incertain environments or applications, the transmitter and receivercomponents of the transducer 12 are physically separated, with thereceiver being located on an opposite side of the object being tested.Hence, the term “echo” as used herein also pertains and encompassesembodiments where the so-called echo signal passes through the objectbeing tested.

In the preceding description, the invention that has been describedexclusively with respect to embodiments wherein flaw detection iscarried out with a single transducer element operating exclusively underthe echo principle and/or by reference to a transmitter/receiver pairwhich handle ultrasound waves that pass through a material. However, itshould be noted the present invention is equally applicable to flawdetection instruments that use an array of transducer elements, such asan ultrasonic phased array probe. As is the case with a single elementultrasonic transducer, the response signal for each transducer elementof the phased array ultrasonic probe used for reception is provided tothe input of a receiver channel for conditioning and subsequentdigitization by an analog to digital converter. In other words, thereference in the claims to a “transducer”—in the singular—is deemed topertain to an ultrasonic phased array type of a probe as well. Sucharrays of transducers are deemed to be either identical or at leastequivalent to a single element transducer. The structure of suchultrasonic phased array devices is described or referenced in U.S. Pat.Nos. 4,497,210 and 6,789,427, the contents of which patents areincorporated herein by reference.

Although the present invention has been described in relation toparticular embodiments thereof, many other variations and modificationsand other uses will become apparent to those skilled in the art. It ispreferred, therefore, that the present invention be limited not by thespecific disclosure herein, but only by the appended claims.

1. An object inspection system, comprising: a transmit and receivesection to generate a test signal and to receive a responsive echosignal, a transducer that converts the test signal to an ultrasonicsignal, applies the ultrasonic signal to a target object to be tested,receives an ultrasonic echo signal and produces the echo signal for thetransmit and receive section; a signal processing circuit coupled withthe transmit and receive section for receiving and processing the echosignal, the signal processing circuit including at least two signalprocessing channels, each channel including a respective preamplifierthat scales the echo signal to a different degree; an analog to digitalconverter coupled to the preamplifiers via amplification circuits, in amanner that passes to the analog to digital converter only anon-saturated output of at least one of the preamplifiers; and anautomatic gain control circuit which is coupled to outputs of thepreamplifiers and is configured to provide signals to an acquisitionlogic circuit that calculates a system gain based on an output of theanalog to digital converter and said automatic gain control circuitprovides gain settings to the amplification circuits to assure thatnon-saturated amplification circuit outputs are provided to the analogto digital converter.
 2. The system of claim 1, wherein said acquisitionlogic circuit is coupled to and receives the output from the analog todigital converter and an additional output which is derived from theautomatic gain control circuit.
 3. The system of claim 2, wherein theadditional output is produced by at least one analog to digitalconverter provided between the acquisition logic circuit and theautomatic gain control circuit.
 4. The system of claim 1, wherein theamplification circuits comprise at least one multiplier circuit.